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High Performance PM Synchronous Motor Drive High Performance PM Synchronous Motor Drive for an Electrical Scooter Bianchi N., Bolognani S., Zigliotto M. Department of Electrical Engineering, University of Padova Via Gradenigo, 6a, 3513 I Padova, Italy fax: +39 .049 . 827 7599, e-mail bolognan@dei...

High Performance PM Synchronous Motor Drive
High Performance PM Synchronous Motor Drive for an Electrical Scooter Bianchi N., Bolognani S., Zigliotto M. Department of Electrical Engineering, University of Padova Via Gradenigo, 6a, 3513 I Padova, Italy fax: +39 .049 . 827 7599, e-mail bolognan@dei.unipd.it Abstract - In this paper, the design of a full electrical system for an electrical scooter is presented. The paper explains the design choices that have been done to satisfy the typical main requirements for an electrical vehicle. According to them, an interior permanent magnet synchronous motor has been incorporated in a drive able to control the motor both in the constant torque (constant flux) and in the constant volt-ampere (flux-weakening) regions. Experimental results carried out on a system prototype are also included in the final paper. an anisotropic rotor, with a fairly reduced PM flux. A commercial 24-slot induction motor stator is used, with a single-layer winding. Fig.1 shows two pole-pitches of the motor. x barrier I. INTRODUCTION Increasing awareness of air quality and interest in innovative vehicles stimulate research activity to improve the propulsion systems by reducing the vehicle .emissions. In recent years, considerable developments have been obtained in the area of electric and hybrid vehicles, whose use solves the environmental problems caused by the use of internal combustion engine vehicles. Moreover, the torque generated by the electric motor can be appropriately controlled so that the vehicle stability and safety result greatly improved. In this paper, the design of a full electrical system for an electrical scooter is presented. The paper explains the design choices that have been done in order to satisfy the typical main requirements for an electrical vehicle, namely: wide flux-weakening (FW) region, to minimise the drive power rating in spite of a wide speed range; high-efficiency to increase the vehicle autonomy; high torque-to-ampere and torque-to-volume ratios to minimise the inverter cost and the drive overall dimensions; simple manufacture for a cost-effective end product. Several considerations are given explaining the design choices for both motor and control. Experimental tests validate the theoretical results. 11. SYNCHRONOUS MOTOR DESIGN The aforementioned goals can be achieved by the use of a synchronous Interior Permanent Magnet (IPM) motor drive, even in case of FW, as widely reported in the specialised literature [ 1-51. The most suitable motor structure considers Fig.1- Schematic ofthe interior PM motor A. Preliminary design The design of the IPM motor for FW operations is normally a two-step procedure. Firstly, the design requires to determine a set of motor parameters that satisfies the required performance in the whole operating range. The parameters used during this first rough design step are usually normalised [6-81, as reported in Appendix. In the following, normalised parameters will be written in italic with small letters. Secondly, the actual motor sizing is developed from the normalised parameters issued by the first step. The procedure has been already described in [9- 101, and it will be here omitted. On the contrary, some considerations on the first design step are reported in the following. It can be fbrther subdivided into three phases, listed hereafter: The saliency ratio €,=ldld is first selected, in accordance with typical values of the adopted rotor structure (Fig. 1). In the present work, the range is between 3 and 5, and an initial guess value 5=4 has been chosen. As clarified below, further coefficients will be futed to consider the saturation effect. Starting ffom the specifications on FW speed & and torque z j j , the suitable PM flux linkage A,,, is chosen. From 5 and ;Im the corresponding normalised q and d axis inductance lq, Id and the rated stator current iiy are computed. The design parameters have to comply with torque and voltage specifications, as well as with maximum torque-to- current ratio relationship [lo]. Even if there is a certain 0-7803-6401-5/00/$10.00 0 2000 IEEE 1901 degree of freedom in the choice, it is important to recall that different combinations of iq, iN and Am do not yield to the same FW performance. Furthermore, the effects of motor non-linearity, due to iron saturation, has to be taken into account [ll-121. For the actual application, a constant power for a speed range equal to three, i.e. power pfiu"l at 9 = 3 (thus r/,.=l>.33 at %=3) and, of course, f b = l at cwl were required. A simple piecewise linear model, that deals with the q-axis saturation, has been introduced and verified by the authors in [13]. It considers a constant PM flux linkage and a constant d-axis inductance, together with two values of the q-axis inductance: the unsaturated lqu and the saturated (differential) fqs . In particular, it has been demonstrated that a saturated motor has to be designed with higher PM flux linkage and lower inductance than the corresponding unsarurated motor. The model has been used in [14] to obtain the normalised motor parameters. F i g 2 reports the FW torque rfi and power ( z j j 9 ) at FW speed ~ q i ~ 3 as functions of the PM flux linkage and ratios <=iqJld and (s=iqJfqs. It is worth to highlight that cs is related to the magnetic characteristics of the q-axis path and that (,=1 refers to an unsaturated motor. Typical values for the selected configuration range from 5,=3 to 7 and here an initial guess 5,=5 has been chosen, as reported in Fig.2. The curves shown also depend on the ratio o,=iq.diq, where iqs is roughly the current at which saturation begins while iq is the rated base current. It is thus evident that oq indicates the saturation level: a unity value refers to an unsaturated motor, and hence a poor exploitation of the magnetic paths, while lower values indicate an higher saturation level [14], that can yield higher iron losses. Winding thermal limits also comes into play if too low oq are selected. Typical trade-off values are from 1 down to 0.4. I;. FW power and FW torque ?li" 1 s 1.35 1 2 1 os 0 9 0.75 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 A m Fig.2 - FW performances, at different saturation levels The experimental prototype developed for this work has been indicated with a black mark on Fig2 and Fig.3. Obviously, it is the consequence not only of the given specifications, but also of the intermediate design choices of & and or. Fig.2 returns a value of the PM flux linkage that can be entered into other two curves obtained from the same simplified model [I41 and reported in Fig.3 (a) and (b). unsaturated q-axis inductance 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 A m rated current 0 0.1 0.2 0.3 0.4 0.5 0.6 07 0.8 0.9 1 A m Fig3 - (a) Normalised q-axis inductance and (b) stator current These characteristics are then exploited to get the values of I,, and IN. Fig.4 resumes the design steps outlined above. Fig.4 - Choice of the normalised IPM motor parameters 111. ANALYSIS AND EXPERIMENTAL MEASUREMENTS ON THE MOTOR PROTOTYPE This Section presents some results obtained from Finite Element (FE) analysis on the motor prototype. They are then validated by means of experimental measurements. 80 70 60 50 40 current 30 20 10 q-axis (A) -80 -70 -60 -50 -40 -30 -20 -10 0 d-axis current (A) Fig.5 - Simulated and experimental results: constant torque loci 1902 Wlth reference to the designed IPM motor, Fig.5 shows the constant-torque loci in the plane (Id-L,), solid l i e . Three current-limit (peak value) circles are drawn in dashed line Black dots in the figure indicate the measured values of motor torque, at the corresponding current operating point. F ig5 highlights that the measured results are in agreement with the simulated ones. I t I S well known that a higher efficiency can be achieved in each drive working condition by feeding the motor according to the maximum torque-to-current ratio strategy. The effective match between FE analysis and experiments enables the use of the former to track the maximum torque- to-current ratio trajectory shown with bold line in Fig.5. AS a result, Fig.6 reports the d-axis and q-axis currents as functions of the stator current magnitude is=1iJ, particularised for the actual parameters of the motor prototype. Y Y 0 IO 20 30 40 50 60 70 80 90 current magnitude (A) I-ig.6 - The d- and q-axis current components for maximum torque-to-current ratio (actual base point of operations, not normalised) Iv. DRIVE STRUCTURE DESIGN The application of the IPM motor drive to an electrical scooter allows enlarged speed range in the constant power conditions, with an inverter size lower than in a conventional flux-oriented induction motor drive [ 1,2]. AS a counterpart, the presence of both reluctance and PM torque requires involved control strategies, to guarantee effective drive operations at all operating conditions. The block schematic of the complete drive structure is sketched in Fig.7. The throttle knob position (tkp*) is here considered as a speed reference, and it is compared with the actual mechanical speed a,, as shown in Fig.7. A light pure proportional control P then delivers the stator current reference space vector is*. The speed loop aims to emulate the behaviour of a fuel engine scooter, in which there is a steady state speed error proportional to the load, i.e. the longitudinal slope of the road SVM inverter Id Fig.7 - IPM motor drive block schematic The main issues for an efficient control are a proper current settings delivery and the avoidance of any inverter voltage saturation. Both aspects will be examined carefully in the next sections. A. The current reference generator The amplitude of stator current i, is directly linked to the torque production. Nevertheless, there are infinite couples id, i, that give the same vector is, each producing a different torque value. In the constant torque operating range various profiles in the id, i, plane can be selected, depending on the optimisation objective of the specific application [ 151. In this work the maximum torque per ampere strategy has been adopted, due to its simplicity and operational significance [ 161. For a given id, i, pair and under the hypothesis of system linearity the electromagnetic torque is given by T =-pi,[A,-(L,-Ld)id] 3 . 2 If the direct current reference id* is expressed as function of i,'and the current reference amplitude is* it is possible to determine the value of the reference i,* that maximises the torque, by imposing that gives in turn 1903 in which resistive drops have been neglected. It is then is the electrical speed (pw,) and the stator i* (1 = sign(r) (4) (7 ) where ida=Am/(Ld'Lq). It is soon evident that, in presence of deep saturation of the q-path, expressions (2) and (4) are only rough and ready approximations of the correct d-q current references. Their use implies less efficient current settings for a given torque request, with increased energy consumption. In this work, FE method has produced accurate estimations of d-q axes current references as function of stator current amplitude (Fig.6). The analysis was based on the search of the couple id, iq that maximises the torque, now calculated by i; = me It iS worth to note that the theoretical Value (7) is function of either current and voltage vector amplitude and of the motor speed as well. The use of (7) as current reference generator requires the knowledge of the electrical speed of in which the FW has to be triggered, given by where hd'hd(id) and hq=hq(iq). The d-q axis cross- coupling, i.e. the effect of each current on the other axis, is here neglected, since it yields a few percents error in the torque computed by (5). The torque value ( 5 ) compared with (1) is more accurate since the former holds also in presence of magnetic non-linearity. For each quantified is* value, the id, iq references of Fig.6 have been stored in a look-up table, that can be used directly as optimal current reference generator in the constant torque region (Fig.7). The extended torque-speed operation of the IPM motor can be reached by increasing negative values of the direct current component, that in turn produces a d-axis flux linkage opposite to the PM flux linkage (FW control) [7]. The generation of the additional FW direct current reference id; is not a trivial task, since it results from a trade-off between the best exploitation of the constant- torque range and a prompt voltage fimitation prevention. That is, an early or too strong FW action would reduce for no reason the available maximum torque, while with a delayed or too weak action the voltage limits would be overcome, with subsequent cross-coupling effects on the current regulation loops and possible instability. A common way to derive the adequate current references for FW is based on the calculation of the intersection between the actual current circle and the voltage limit ellipse in the id-& plane [17]. For a given current is* and a rated amplitude Ulim of voltage vector the id*, i,* couple can be derived from (2) and the steady state voltage balance equation Since current measurements are available, it is possible, in principle, to use the FE data to obtain the exact values of the inductances Ld and L,, also considering the saturation effects. These values can be used to decide by (8) whether the FW strategy (7) has to be started or not. It is clearly a cumbersome procedure, that takes up a considerable amount of processing time. It is also noteworthy that during transient operation below wfthe voltage reference vector ua '+jup* could still exceed the available inverter voltage, and the actual currents could not follow the references, with performance degradation [ 151. All the above indicates that automatic FW triggering and current setting modification would be a great improvement, especially in battery-energised applications in which every waste must be carehlly avoided. An effective solution is the realisation of an outer voltage loop, (Fig.7), which delivers a FW component id< that is added to the actual id* when the SVM (space vector modulated) inverter voltage limit is approached. The quadrature current reference i,' is now limited by (9) This solution was first proposed by 1171. Here it is presented an improved version with some innovative details that sensibly improve the system performance under particularly stressing operational conditions. 1904 B. The voltage control loop First of all, an accurate linearisation around the voltage limit working point has confirmed the experimental evidence that the proportional part greatly reduce the stability of the voltage loop, and therefore a pure integral action is to be preferred. The design now focuses on the choice of KIu and Kv, that fixes the reference voltage limit of the loop (Fig.7). The maximum pure sinusoidal voltage vector that can be produced for a given dc link is O.577*Udc. Of course, K, must be less than 0.577, and in this work a value K,=0.54 has been used. It must be mentioned that exceedingly low values trigger the FW in advance, as shown in case (b) of Fig.8. 6 ' t 5 . 4 . 3 - 2 - 1 - OL I 0 100 200 300 400 500 600 w , (radls) Fig.8 - IPM drive torque-speed behaviour. (a) without FW (b) Kv=0.5 (c) Kv=0.54 Fig.8 reports the actual drive torque-speed behaviour. As expected, in absence of any FW action the voltage limit is soon reached and the delivered torque falls almost immediately as the base speed is exceeded, while an accurate choice of K, yields the full exploitation of the constant torque range as reported in case (c). It has also been found that by using a pure integral action, the choice of the gain constant Ki, is not critical. In complete saturation condition, the actual voltage vector traces the hexagonal boundary. The maximum (negative) input error is given by the difference between the actual voltage amplitude and the fixed limit Kv*Udc (Fig.7). As the FW action starts by producing an increasing idf Component, the error decreases to zero. As a simple tule of thumb, it is reasonable to choose K;, so thal in presence of the maximum input error the rated FW component i&;N is delivered within a fixed time, let's say 30 t imes the id current loop rise time tri. Since the amplitude of ihe fundamental voltage in case of hexagonal saturation is 0.6056 u d c [IS], K;, can be calculated as K. = Idf ,N 30t,; (0.6056- K,)Udc IU Several simulation results have confirmed that if K,, is chosen sensibly smaller than (10) the voltage loop is no more able to follow the transient voltage produced by the current control and inverter saturation occurs. Conversely, if the voltage loop is too fast, instability due to system delays have been experienced. Another crucial design item is the maximum negative FW current whose value has to be chosen according to the actual working point by where IN is the rated stator current and idt is the direct current that the current reference generator would give in absence of voltage limitation. Actually, if a fixed limitation would be chosen instead, instability may arise at load detaching during deep FW operations. Fig9 illustrates the principle. -\ voltage current limit cimstant limit 'df,N mdfN\;/ idt 1 Fig.9 - Voltage loop output limitation design Let's consider point A the base working point, and assume that a constant maximum FW current idf,mln=idf,N is selected. If the IPM is accelerated up to two times the base speed, let point B be the actual intersection between the voltage limit and the stator current required by the speed loop, At steady state, the inertia torque component drops off sharply and the stator current request also reduces (small circle in Fig.9). Due to the given idf,min limitation, point C becomes the reference working point, and since it lays outside the voltage limit ellipse, the current control could become unstable. Conversely, if (10) is used the integral loop reacts and the final stable point D can be reached. Fig.10 shows the different FW components generated with fixed output limitation (case 1) and using (1 0) (case 2, dashed line). 1905 -do1 -60 b 1 2 3 4 5 k-*O0 -80 f (SI Fig.10 - FW action at load torque detaching Fig.11 and Fig.12 show the effects on different FW actions on current control. In particular, with fixed limitation (case I ) both control loop loose current control, and oscillations arise around the final steady state working point. Perfect current control is maintained using the proposed method (case 2). -80 0 1 2 3 4 5 6 Fig. I I - id current control at load torque detaching J t (SI 30. 20 10. 0 . -'Ob 1 2 3 4 5 L t (SI Fig. 12 - i 4 current control at load torque detaching The calculation of the actual stator voltage is based on voltage references (Fig.7). To avoid the imprecision essentially due to dead times and voltage drops on switching devices, an appropriate dead times compensation (DT cmp) algorithm [ 191 has been implemented. Finally, the voltage saturation can also be sensed simply by monitoring the period To of application of the zero-voltage vector, as delivered by the SVM algorithm. C. Management of current regulators saturation levels In every modern AC drive the current control plays a crucial rule and can be fully considered a distinguished feature of the drive itself. Actually, a proper current control can be achieved in the synchronous d-q coordinates system by conventional PI contr
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