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简介:本文档为《A New Control Technique for Achieving Wide Constant Power Speedpdf》,可适用于电子通讯领域,主题内容包含ANewControlTechniqueforAchievingWideConstantPowerSpeedOperationwithanInter符等。

A New Control Technique for Achieving Wide Constant Power Speed Operation with an Interior PM Alternator Machine Jackson Wai Thomas M. Jahns University of Wisconsin-Madison 1415 Engineering Drive Madison, WI 53706 Ph: 608-265-3821 Fax: 608-262-5559 E-Mail: waij@cae.wisc.edu University of Wisconsin-Madison 1415 Engineering Drive Madison, WI 53706 Ph: 608-262-5702 Fax: 608-262-5559 E-Mail: jahns@engr.wisc.edu Abstract-A new flux weakening control algorithm is presented for achieving wide constant power operation with an interior permanent magnet (IPM) synchronous machine operating as an alternator. The algorithm is designed to initiate flux weakening only when necessary by recognizing the threshold conditions for current regulator saturation, making the algorithm inherently adaptive to changes in the inverter bus voltage. A 6kW 42V automotive integrated starter/alternator (ISA) using a direct-drive IPM machine serves as the target application for this development. Simulation results show that the new algorithm is capable of delivering very good static and dynamic bus regulation characteristics over a 10:1 engine operating speed range. Tests are presently under way to confirm these performance predictions using a prototype IPM starter/alternator system. Keywords: Permanent magnet machines, AC motor drives, road vehicle power systems, power control, synchronous generators I. INTRODUCTION With proper design, an interior permanent magnet (IPM) synchronous machine can achieve constant power operation over a wide speed range in both the motoring and generating regimes [1]. As a result, IPM machines are attracting increasing attention for a variety of applications ranging from electric vehicle propulsion [2] to machine tools [3]. While the design of an IPM machine for extended speed operation poses several technical challenges, the success of the complete drive system depends on the availability of an appropriate control algorithm that can extract the full performance capabilities from the machine. In particular, special flux weakening control techniques are necessary to achieve constant power operation over a wide speed range. Previous work reported in the literature has generally focused on extended speed range applications for the IPM machine operating as a motor [4-6]. This paper presents a new control algorithm designed to extract near-constant power from an IPM machine over a wide speed range when it is operating as an alternator. The application that has motivated this work is a direct- drive integrated starter/alternator (ISA) for an automotive vehicle [7]. The development of many new electrically- S/A rotor acts as flywheel Torque converter inside rotor hub Flywheel, alternator, starter, pulley, & possibly a belt eliminated New Parts Eliminated Parts Fig. 1 Integrated starter/alternator (ISA) configuration. powered accessories combined with wide interest in mild hybrid concepts using electric machines for low speed acceleration has drawn significant attention to ISA systems [8]. As illustrated in Fig. 1, the new ISA replaces the conventional engine flywheel and is driven directly by the engine crankshaft without gearing. The machine is designed to operate as both the engine starter and alternator, replacing the two separate machines that perform these functions in conventional vehicles, along with their associated pulleys, belts, and gears. Performance specifications developed in consultation with several automotive manufacturers call for the ISA to supply 150Nm of starting torque at low speeds. In addition, the ISA must deliver 42V power to the automotive accessories over a 10:1 speed range extending from 4KW at 600rpm to 6KW at 6000rpm. It is this latter requirement that motivated selection of the IPM machine for further development as an attractive candidate for this application. Significant progress has been made towards development of an IPM machine that is capable of meeting these challenging application requirements [9]. This effort has culminated in construction of a prototype version of a 12- pole IPM machine designed for this direct-drive ISA application. Figure 2 shows the stator and rotor of this machine, illustrating its use of two magnet cavities per pole with saturating magnetic bridges that link the iron pole pieces into unitary rotor laminations. 0-7803-7116-X/01/$10.00 (C) 2001 IEEE 807 Fig. 2 Rotor and stator assemblies of prototype 6 kW IPM machine for ISA application. II. CONTROL TECHNIQUE PRESENTATION A. IPM Machine Operating Envelope The principles for achieving wide constant power operation using IPM synchronous machines have been established in the technical literature [4,10]. The machine equations governing the dynamic operation of the IPM machine in the rotor dq reference frame are summarized below. Note that the d-axis is aligned with the rotor permanent magnet flux. vd = rs id + plds - wrlqs (1) vq = rs iq + plqs + wrlds (2) lds = Lds id + fpm (3) lqs = Lqs iq (4) where id and iq are the d- and q-axis stator currents [A] vd and vq are the d- and q-axis stator voltages [V] lds and lqs are the d- and q-axis stator flux linkages [Wb] Lds and Lqs are the d- and q-axis stator inductances [H] (Note that Lqs> Lds in an IPM machine) fpm is the d-axis permanent magnet flux linkage [Wb] wr is the rotor rotation frequency [elec. rad/s] Magnetic saturation has a significant impact on the operating characteristics of an IPM machine. Experience with the automotive ISA machine [9,11] has indicated that saturation can be modeled quite accurately by making Lqs a function of the q-axis stator current iq while treating Lds as a constant. Cross-coupling saturation effects [12] between the q- and d-axes are not significant in this machine and are not included in this model. Equations (1) and (2) can be rewritten using (3) and (4) plus chain differentiation to reflect the fact that Lqs is a function of iq: vd = rs id + Lds pid - wr Lqs iq s (5) vq = rs iq + Lqs' piq + wr( Lds id + fpm) (6) where Lqs' = q qs i L iq + Lqs (7) Previous work [4,10] has demonstrated the value of the dq current plane for evaluating the extended speed operating characteristics of IPM machines. More specifically, such plots provide a means of visually depicting the constraints imposed by the inverter voltage and current limits and their interactions. Figure 3 presents the dq current plane plot for the ISA machine described above whose electrical parameters are provided in the Appendix. The Appendix includes a numerical expression for Lqs as a function iq that has been fit to machine's inductance characteristics. The current limit manifests itself in Fig. 3 as a circle of amplitude 326A (peak) centered at the origin, while the voltage limit takes the form of a family of nested ellipses all centered at the IPM machine's characteristic current value, -fpm/Lds. The radii of the ellipses vary inversely with the rotor speed. These ellipses are distorted in the vertical (iq) direction because of Lqs saturation effects, and their longest diameters exhibit a noticeable counter-clockwise tilt from the horizontal (id) axis because of stator resistance effects. At any given speed, the IPM machine can operate at any combination of iq and id values that falls within the overlapping area of the current limit circle and the voltage limit ellipse associated with that speed. -800 -600 -400 -200 0 200 400 600 -600 -400 -200 0 200 400 600 id [A] iq [A ] Current Limit Circle Imax Voltage Ell ipses Max T/A without saturation Max T/A with Lqs saturation effect Motoring Generating -ff P M/Lds ww r increasing Generat ing Mode Envelope Trajectory Fig. 3 Plot of id-iq current plane for the 6kW IPM machine showing current limit circle, voltage limit ellipses, and current vector trajectories. 808 Figure 3 identifies the maximum torque-per-Amp current vector trajectory for the ISA machine in the second quadrant motoring regime for starting operation, with and without the effects of magnetic saturation. The trajectory without saturation forms an angle with the negative –id axis that exceeds 45 degrees, while the angle of the corresponding trajectory with saturation is less than 45 degrees. During generating operation of the IPM machine at elevated speed, the maximum output power point follows the periphery of the current limit circle towards the negative id axis as indicated in Fig. 3. This motion is forced by the increasing speed that progressively shrinks the voltage limit ellipse, preventing the machine from operating in the vicinity of the maximum torque-per-Amp trajectory (for generating) identified by a dashed line in Fig. 3. The amplitude of the ISA machine's characteristic current, fpm/Lds (=135A) is less than the inverter current limit (326A). This may seem surprising since optimum extended speed operation occurs when the value of fpm/Lds equals the inverter current limit [1]. However, the other ISA system specifications, including a limit on the machine’s maximum back-emf amplitude, result in this lower value of characteristic current. Given this situation, the maximum generating output power trajectory eventually separates from the current limit circle as the speed increases, following a trajectory that leads to the -fpm/Lds characteristic current point on the negative id axis at infinite speed. It is important to note that the illustrated generating mode trajectory in Fig. 3 for speeds above the corner point actually represents an optimistic outer limit for the current vector locus that can only be approached but never quite reached for an actual current regulated drive. This is true because the outer boundary of the voltage ellipse at any speed corresponds to six-step voltage operation, representing a condition in which the current regulator loops are completely saturated. Since the current regulator loses control of the instantaneous machine phase currents under such conditions [4], the current vector command must be continually adjusted so that it always resides safely inside the voltage ellipse. However, it is desirable to approach the ellipse as closely as possible under heavy load conditions in order to deliver maximum power from the IPM machine, taking full advantage of the available inverter dc bus voltage. With this observation in mind, it is clear that angle between the commanded current vector and the negative d-axis in Fig. 3 must be reduced as the shrinking voltage ellipse progressively intrudes on the current limit circle for speeds above the corner point. This control action, illustrated in Fig. 4, provides the basis for the new generating mode control algorithm described in the next section. IS1 IS2 q2 q1 - Iq - Id Flux Weakening Action Fig. 4 Plot of current vectors illustrating flux weakening action during generating mode operation. Starter/ Alternator Machine Shaft Angle Transducer T e * i q 42V + - DC Link Capacitor Vector Rotator V d c V dc V dc T Start (For Starting) V dc * (For Generating) + - P I S/A Control Module id V q e* V d e* V q s* V d s* qqr V dc Gate Signals Space Vector Modulator Vector Rotator d/dt ww r qq r qq r Fig. 5 Block diagram of ISA controller configuration. B. Controller Configuration A top-level block diagram of the ISA controller is provided in Fig. 5. This figure includes many of the features that typically appear in high-performance vector control configurations utilizing synchronous-frame current regulation. That is, phase current information is combined with rotor position feedback to develop measurements of the instantaneous id and iq that are fed to the current regulator imbedded in the block labeled S/A Control Module. Another input to this module is the torque command Te* that is derived from either the starting torque command or the bus voltage regulator, depending on the system's operating mode. The outputs of this module are the voltage commands vde* and vqe* in the synchronous reference frame. These voltage commands are then converted back to the stationary reference frame using a second vector rotator before being applied to a space vector modulator. This modulator develops the inverter switching commands, making use of the measured bus voltage vdc to insure a smooth transition from PWM operation into saturated six-step voltage operation under all bus voltage conditions [13]. 809 Figure 6 shows the detailed structure of the S/A Control Module that appears as a block in Fig. 5. The control components enclosed by the inner dashed-line box in the lower half of Fig. 6 are responsible for flux weakening operation and are only are activated at elevated speeds when the current vector approaches the prevailing voltage limit ellipse at any given speed. Under all other conditions, such as low-speed starting operation, the flux-weakening controller is essentially dormant, having no effect on the control action of the Fig. 6 S/A Control Module. The control elements lying across the top half of Fig. 7 dominate its response under such conditions, as described below. Starting Operation The vector control principles for controlling the ISA machine to deliver high starting torque (up to 150 Nm) as a motor are well established in the literature [14] and will be summarized only briefly here. Since starting mode operation occurs only at low speeds where the voltage limit ellipse is not a factor, the maximum torque-per-Amp trajectory in the second quadrant provides the preferred operating points for motoring torque production. The maximum available starting torque from the ISA machine occurs at the point in the second quadrant where the maximum torque-per-Amp trajectory intersects the current limit circle. The S/A Control Module in Fig. 6 executes this starting mode control by converting a starting torque command Te* into the corresponding values of d- and q-axis current commands, idm and iqm, using the function blocks f1 and f2 and the polar-to-rectangular conversion block. These f- functions are pre-programmed with the maximum torque- per-Amp trajectory that matches the IPM machine's characteristics. The two f-functions develop the desired current vector trajectory in polar coordinates as current amplitude |I|* and angle q* commands (see Fig. 5) for convenient compatibility with the flux weakening control algorithm that will be described in the next section. The q* angle command passes through to the polar-to-rectangular converter unaffected by the flux weakening control algorithm at low speeds during starting. The upper right side of Fig. 6 represents the stator current regulator in the synchronous reference frame. There are many different ways in which such a regulator can be implemented. The particular implementation shown in Fig. 6 applies feedforward decoupling compensation to the output voltage commands using the machine's steady-state voltage relations, so that: vqss = rs iqm + wr (Lds idm + fPM) (9) vdss = rs idm - wr (Lqs iqm) (10) Polar To Rect Voltage Compensation Block iq m id + - + - + + + + vd ss vq e* qq *' | I |*T e * Modulation Index Calculator+ - Mth bb ** Flux Weakening Control Module S/A Control Modlule A S vdc G G Integrator Maximum Torque/Amp Functions 00 << b b ** << 1 1 id m iq DD iq DD id vq ss vd e* vq e' vd e' f2(T) f1(T) vq e* vd e* qq ** ww r M Fig. 6 Block diagram of S/A Control Module. Generating Operation Development of an effective flux-weakening control algorithm provides the key to achieving extended constant- power operation to meet the demanding generating mode requirements of the automotive ISA application. The new algorithm implemented in this system works by directly reducing the angle q of the commanded current vector with respect to the negative d-axis as illustrated earlier in Fig. 4. The control elements that implement this flux weakening action are enclosed in the sub-block labeled Flux Weakening Control Module in the lower half of Fig. 6. This module executes the current vector angle reduction by multiplying the angle command q* from maximum torque- per-Amp function f2 by a scaling factor b* that varies between 0 and 1. According to this approach, only the angle of the current vector, and not its amplitude, is directly controlled by the flux weakening algorithm. As long as the value of b* is 1, the Flux Weakening Control Module exerts no influence on the current vector trajectory defined by the maximum torque-per-Amp trajectory. The value of b* is determined by how closely the required stator voltage approaches the saturated six-step voltage value at any instant in time. Since the ISA drive is designed for high frequency PWM operation, the instantaneous PWM modulation index, M, can be used as a convenient and sensitive indicator of the current regulator’s proximity to total saturation (i.e., M = 1). The modulation index value can be conveniently calculated using the synchronous frame dq voltage commands, vqse* and vdse*, as follows: M = dc 2*e qs 2*e ds V2 )v()v( p + (11) where Vdc is the measured value of the dc link voltage at the converter input terminals. 810 Any excess value of M above a preset threshold level, Mth (less than but close to 1) is integrated to reduce the value of b* (0<b*<1) that scales the current vector command angle according to q*’ = b* q* . This feedback loop achieves the desired flux weakening action in Fig. 4 by swinging the current vector in towards the negative d-axis in order to retain it within the prevailing voltage ellipse at that time instant. By this action, the value of the modulation index naturally decreases as the current vector operating point moves away from the voltage ellipse boundary, establishing a stable operating point loop in the flux weakening regime. When the torque command and/or rotor speed drop sufficiently so that flux weakening is no longer required, the feedback loop naturally responds accordingly to reduce its effect on the current vector angle q. That is, the value of b* quickly integrates back to 1 when the modulation index drops below its threshold value M*. Thus, the feedback loop becomes passive with no effect on the vector control operation until flux weakening action is required again. Use of the measured value of dc link voltage Vdc in the modulation index calculation (11) makes it possible for the controller to adapt immediately to any changes in the bus voltage. This feature is important in applications such as an automotive ISA where the dc bus voltage can be expected to change significantly depending on the bus loading and the vehicle battery state of charge. On the other hand, this voltage measurement can be replaced by a constant value in applications where the dc bus voltage is known and well regulated within a narrow range, eliminating the need for the bus voltage sensor. The flux weakening algorithm introduced above is naturally bipolar, working as well for motoring operation as for generating where it is primarily needed in the ISA application. The resulting motoring torque envelope along the voltage ellipse boundary for the IPM machine in this ISA application is shown in Fig. 7. This figure confirms that the required 150 Nm for starting will be available at low speeds during operation on the maximum torque-per- Amp trajectory. IV. CONTROLLER PERFORMANCE A. Drive System Simulation Steady-state and dynamic performance characteristics of the complete ISA drive system including the machine, PWM inverter and the controller have been investigated using computer simulation. Matlab/SimulinkTM was chosen as the preferred simulation software tool for carrying out this analysis. The switching behavior of the inverter is modeled using six ideal switches. The IPM machine is modeled in the dq synchronous

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