IEEE Industry Applications Society
Annual Meeting
New Orleans, Louisiana, October 5-8, 1997
The Quasi-Distributed Gap Technique for Planar Inductors:
Design Guidelines
Jiankun Hu Charles R. Sullivan
Thayer School of Engineering
Dartmouth College
Hanover, NH 03755, USA
Phone: (603)643-2851 Fax: (603)643-3856 E-Mail: Charles.Sullivan@dartmouth.edu
URL: http://thayer.dartmouth.edu/inductor
Abstract -- The use of low-permeability magnetic material to
form a uniformly distributed gap can facilitate the design of
low-ac-resistance planar inductors. Alternatively, a quasi-
distributed gap, which employs multiple small gaps to
approximate a uniformly distributed gap, can effect the same
result. Finite-element simulations are used to
systematically analyze the design of quasi-distributed gap
inductors. It is shown that a good approximation of a
distributed gap is realized if the ratio of gap pitch to spacing
between gap and conductor is less than four, or if the gap
pitch is comparable to a skin depth or smaller. For a
practical range of gap lengths, the gap length has little effect,
but large gaps can reduce ac resistance. An analytic
expression, which closely approximates ac resistance factor
for a wide range of designs, is developed.
I. INTRODUCTION
Conductor losses in high-frequency magnetic
components are strongly influenced by the magnetic field
configuration and distribution in the winding area. For
transformers, it is relatively straightforward to understand and
to control the field and the resulting losses. However, in
inductor designs, the field configuration is influenced by the
geometry and position of the gap as well as the conductors.
Avoiding excessive losses can be challenging.
Inductors in planar configurations are often desirable,
either because of packaging constraints, or because of the
fabrication technology. A planar configuration has the
potential for particularly high ac conductor losses. This
problem has been addressed extensively in the literature, e.g.
[1, 2, 3, 4, 5, 6]. One of the most elegant solutions is to use
a low-permeability material to effect a distributed gap across
the top of the planar conductor [2], as shown in Fig. 1.
Since high-performance, low-permeability power materials are
often not readily available, the “quasi-distributed gap” 11
11 Denoted as a “discretely distributed gap” in [9]
(shown in Fig. 2), which uses multiple small gaps to
approximate a lower-permeability material, has been proposed
as an alternative to the true distributed gap [6, 7, 4, 8, 9, 10].
Figure 1. Distributed-gap inductor: use of a low-permeability
material to achieve low ac resistance.
Figure 2. Quasi-distributed gap inductor using multiple
small gaps to approximate a distributed gap
Although the principle of quasi-distributed gaps is well
established, adequate design rules have not previously been
developed. E.g., it is not immediately clear how many small
gaps are necessary to approximate a distributed gap. For any
given design, it is possible to use finite-element simulations
to calculate losses, and use trial and error to find a design that
works adequately. But this is not an efficient approach in
practical design. It would be preferable to be able to calculate
A) Problem Definition and Simplification
We wish to reduce the problem to the minimum
essential framework, to facilitate computations and conceptual
understanding. To do this, we assume an infinitely wide
quasi-distributed gap inductor. In this infinitely wide strip,
with an infinite number of gaps, each gap is equivalent.22
This means that we can simulate only a single gap, as shown
in Fig. 3, with symmetry boundaries on the left and right.
We can describe this structure in terms of six geometric
parameters as shown in Fig. 3: gap pitch p, gap length g, the
space between the conductor and the gaps s, the thickness of
the core tm (assumed equal for top and bottom core sections),
the thickness of the conductor tcu and the spacing between the
lower core and the conductor sb. We normalize all
these
dimensions in terms of skin depth in the conductor material,
d
p m s
=
× × ×
1
0f , and assume the permeability of the
magnetic material is infinite.
If the quasi-distributed gap approximates a distributed
gap well, the choice of conductor thickness is already well
understood: since current will mainly flow in the top skin
depth, a thickness of one to two skin depths is sufficient to
achieve near-minimum ac resistance. Thicker conductors can
be used, but they will decrease only dc resistance. Thus, we
perform most of our simulations with a conductor two skin
depths thick. Since the field has decayed to near zero at the
bottom of the conductor, the spacing to the back magnetic
material, sb, is not important. The thickness of the core
material is also unimportant for the present purposes, as it is
considered an ideal material. This leaves as parameters only
22 An infinitely wide strip like this is an ill-posed problem, in that the inductance per unit width becomes undefined, but
this is not important to the results here.
g, s and p as shown in Fig. 3. By using a systematic
approach to these variables, and by exploiting the symmetry
of the problem to allow finite-element simulation of a section
of length equal to only one gap pitch33, it is possible to
generate sufficient simulation data to understand the problem
thoroughly.
Figure 3. A section of a quasi-distributed gap used for
simulation. Dimensions are normalized to one skin depth in
the conductor.
II. SIMULATION RESULTS AND ANALYSIS
A) Simulation Results
The results of simulations are plotted Fig. 4, which
shows the ac resistance factor
Fr = Rac / Rdc
for various values of gap pitch p and spacing between gap and
conductor s. To explain the results qualitatively, we first
consider the variations in the pitch. As the gap pitch gets
larger, Fr increases significantly. This is due to the tendency
for current to crowd near the gaps, as shown in Fig. 5. As
the gap pitch gets smaller, the region of current crowding
becomes a larger fraction of the overall width. When adjacent
regions overlap (p equal to one to two skin depths), the
current distribution becomes approximately uniform, as
shown in Fig. 6. The ac resistance factor approaches a
minimum, almost equal to the ac resistance factor with a
distributed gap.
33 One could use the lateral symmetry of the structure in Fig. 3 to further shorten simulation time, but this is not
convenient with the software tool we used.
Figure 4: Simulated ac resistance factor, Fr as a function of
gap pitch p, and the conductor-gap spacing s, both normalized
to skin depth. The gap length was 0.1 (one tenth of a skin
depth) for these simulations; however, the results apply for
any small gap.
Figure 5: Current distribution in the cross-section
(p = 5, s = 1, g = 0.1).
Spacing the gaps away from the conductor can also be
beneficial for decreased ac losses. In this sense, intuition
regarding gap fringing fields is correct. The distance required
for any given ac resistance is affected by gap pitch, p, as can
be seen in Fig. 4. One may think of this as the myopia of
the eddy current losses. If the gaps are far enough away, they
“blur out” and “look” like a single distributed gap.
In the above discussions, the gap length g is fixed. In
order to illustrate the effect of gap length on ac losses, two
typical values of gap pitch and spacing from the region
shown in Fig. 4 are selected. The first one represents the case
where original value of Fr is relatively low, while the second
one represents the case
Figure 6: Current distribution in the cross-section
(p = 1,s = 1, g = 0.1).
Figure 7. Ac resistance factor, Fr as a function of gap
length, g, normalized to one skin depth, for two geometric
configurations defined by the normalized values of p and s
indicated (see Fig. 3).
of a higher original Fr value. The simulation results for
these two cases with a range of gap lengths are shown in Fig.
7.
To explain the results shown in Fig. 7, we return to the
idea of current crowding into the region near the gaps. As the
gap gets bigger than one skin depth, the width of the gap
becomes the dominant factor determining the width of the
region of current crowding. Thus a wider gap can spread the
current out further and decrease losses. This tendency can be
seen by comparing Fig. 8 to Fig. 5.
Although wide gaps can reduce ac losses, this is not
useful in most practical situations. The inductance
requirement will typically constrain the total gap length, g(,
to be a fixed, small fraction of the conductor width w.
Suppose n gaps are distributed within w. In this case, g =
g(/n and p = w/n. To get ac losses near those of a true
distributed-gap inductor, one might think that n should be
small in order to make the gap length larger. However, if n
is reduced, the gap pitch will also become larger. As shown
in Fig. 4, making the gap pitch larger will increase Fr
dramatically.
Figure 8 Current distribution along the cross-section
(p = 5, s = 1, two gaps, g = 3).
For the example shown in Fig. 7(a), changing the gap
length can not improve Fr significantly. In the example
shown in Fig. 7 (b), in order to significantly reduce losses
and achieve Fr of about 2.5, the gap length has to be about
50% of the pitch length. This results in an effective
permeability of the quasi-distributed gap of about 2, too low
for most inductor designs. In practice, gap length g is
typically constrained to be small enough that it has little
effect on the ac resistance.
Fig. 7 also shows that gap length has almost no effect
on ac resistance in the region of small gaps. E.g., in Fig. 7
(b), a factor of 30 change in g (from 0.01( to 0.3(, where ( is
the skin depth in the conductor) results in only about a 1.5%
reduction in Fr . Note that this includes a range of ratios of
spacing to gap length from s/g = 10 to s/g = 0.33,
confirming that a rule-of-thumb based on this ratio is not
useful.
Most of our simulations use a small gap, g = ( / 10,
and the results are applicable to any design with a gap that is
small compared to a skin depth, although very large gaps can
reduce losses somewhat relative to the results we report
below. With small gaps, and fixed conductor thickness, Fr
may be described as a function of just two variables, the
spacing from the gap to the conductor s and the gap pitch p.
It is the size of dimensions relative to skin depth that really
matters; thus, the data in Fig. 4, where the dimensions are
normalized to one skin depth, can be used to determine the ac
resistance for any design. (They directly apply only if the
thickness of the conductor is equal to two skin depths.
Scaling for other thickness of conductor will be discussed
later.)
Both in Fig. 4 and Fig. 7, the minimum ac resistance
factor is about Fr = 1.9. A one-dimensional solution of
Maxwell’s equations yields the following expression for ac
resistance factor with a uniformly distributed gap [11], [12]:
F =
2 t sinh(
2 t
)+sin(
2 t
)
cosh(
2 t
)-cos(
2 t
)
r
cu
cu cu
cu cu
× ×
× ×
×
× ×
d d
d
d d (1),
where tcu is the thickness of the conductor. For
t 2cu = × d ,
Fr = 1.8978, very close to the minimum ac resistance factor
of the simulation results.
B) Analytical Approximation
In order to facilitate design without the need to use
tables or plots of data, an empirical expression to describe Fr
as a function of conductor-gap spacing s and gap pitch p is in
need. Using a numerical least-square fit, we developed the
following expression to approximate Fr(s,p):
F (s,p)
k
(b p )
k p 1.9r
n n 1n
=
-
+
+ × +
- -
, (2)
where n 5.4= , (3)
k
0.95
0.95 1.4 s
=
+ × , (4)
b 3.33 s 2.14= × + . (5)
The Fr values computed by the above expressions are
compared to simulation results in Fig. 9. It can be seen that
this expression approximates the simulation results very
well, with relative error less than 4.5%, and absolute error in
Fr less than 0.08. This expression also remains accurate for
configurations with large s and p, as shown in Fig. 10.
For large s, (3) and (4) become:
k
0.68
s
@
(6)
b 3.33 s@ × (7).
We can rewrite the expression (1) as:
Fr
0.68
(3.33 (
p
s
) )
0.68 (
p
s
) 1.9
n
1
n
=
-
+
+ × +
-
(8).
Figure 9: Comparison between values of Fr computed by
using the approximation expression (2) and values obtained
from finite-element simulations.
If the ratio of
p
s is much greater than 3.33, the equation will
enter a roughly linear region and it can be simplified as
Fr 0.68
p
s
0.36= × -
. In contrast, If the ratio of
p
s is much
smaller than 3.33, the equation will enter a constant region
with Fr = 1.9. These two trends are shown in Fig. 10. In a
practical design, if the space is not critical, we can thus make
the spacing s relatively larger, like about one fourth of a gap
pitch, to lower the ac power losses. From Fig. 10, we also
can conclude that the expressions (1), (2), (3), (4) is still a
quite accurate model of Fr even for large conductor-gap
spacing.
Figure 10: Fr as a function of the ratio p/s with s = 20. The
results are approximately the same for any large value of s.
C) Full Device vs. Periodic Segments
To check the assumption that a complete device with
multiple gaps could be modeled by a single segment of a
periodic structure, we simulated two-dimensional cross
sections of complete devices, as shown in Fig. 11. Some
simulations used a symmetry boundary on one side to
represent a device in which the current flows in a planar loop,
returning in a adjacent repetition of the structure in Fig.11. In
all these simulations, the ac resistance factor was within 2%
of the value obtained from the simulation of a single
segment. We conclude that in typical practical designs, the
losses are accurately modeled by a single segment.
D) Finite Permeability of Core Material
Throughout our simulations we assumed that the
permeability of the core material is almost infinite, which
implies that the reluctance of the core is zero and flux is only
determined by the reluctance of gap. In a real design, if we
consider the effect of the finite reluctance of the core, the
fraction of the MMF drop across the core will be higher,
while the MMF drop across the gaps will be lower. The
MMF drop across the core is similar to the MMF drop in a
true distributed gap, and thus, the ac resistance is reduced. In
order to illustrate this quantitatively, simulations using
finite-permeability core material are performed. For a
configuration of p = 5 and s = 1,
Fr|m ' .= ¥ @ 213 ,
Fr|m ' .= @1000 185 ,
where m ' is the relative permeability of core material. These
results confirm the previous discussion. Note that the ac
resistance factor with m ' 1000= is less than 1.9, the
minimum with a distributed gap as shown in Fig. 1. This is
because with low permeability material underneath the
conductor, we start to use the bottom surface as well as the
top surface to conduct ac current.
Figure 11: A full device with four equally distributed gaps.
E) Thickness of the Conductor
In the discussion above the thickness of the conductor
is fixed to be two times the skin depth. Although increasing
the thickness of the conductor will lower the dc resistance, it
will not improve the ac resistance significantly. Given a
thickness larger than two skin depths, we can use (9) to
estimate Fr.
F (t x) F (t 2)
x
2
r r= @ = ×
(9),
where the thickness of the conductor x is greater than two
skin depths.
III. DESIGN RULES
Based on the above analyses, we can glean the
following design rules. Low ac losses (ac resistance factor
less than 2.5(approaching that of a distributed-gap inductor),
can be obtained if either of the following conditions are met:
The ratio of gap pitch to conductor-gap spacing (p/s) is
less than four, or
The gap pitch p is less than 2.5 times skin depth.
If the gap length g is sufficiently large, it can also reduce the
ac resistance. However g must be a substantial fraction of the
gap pitch p to have tangible effect on ac resistance. This is
unlikely to be practical in most designs.
These results can be applied to designs using a wide
range of fabrication technologies and frequencies. For
example, the results could apply to ferrite-core inductors with
mm-scale dimensions used at high frequencies, or inductors
fabricated with thin-film deposition and photolithography,
with (m-scale dimensions used at MHz frequencies. By using
(2), the approximate Fr value can be found to estimate the
configuration of any design.
IV. CONCLUSION
Simulation of a single segment of a periodic quasi-
distributed gap inductor is adequate to predict the ac resistance.
A set of such simulations produces data that can be used for a
wide range of designs. Approximate analytic formulae that
describe these data accurately have been developed. This has
led to a simple set of design rules that can be used to ensure a
design which will have low ac resistance.
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