LM2731[1]

LM2731 April 29, 2010 0.6/1.6 MHz Boost Converters With 22V Internal FET Switch in SOT-23 General Description The LM2731 switching regulators are current-mode boost converters operating at fixed frequencies of 1.6 MHz (“X” op- tion) and 600 kHz (“Y” option). The use of SOT-23 package, made possible by the minimal power loss of the internal 1.8A switch, and use of small in- ductors and capacitors result in the industry's highest power density. The 22V internal switch makes these solutions per- fect for boosting to voltages up to 20V. These parts have a logic-level shutdown pin that can be used to reduce quiescent current and extend battery life. Protection is provided through cycle-by-cycle current limiting and thermal shutdown. Internal compensation simplifies de- sign and reduces component count. Switch Frequency X Y 1.6 MHz 0.6 MHz Features ■ 22V DMOS FET switch ■ 1.6 MHz (“X”), 0.6 MHz (“Y”) switching frequency ■ Low RDS(ON) DMOS FET ■ Switch current up to 1.8A ■ Wide input voltage range (2.7V–14V) ■ Low shutdown current (<1 µA) ■ 5-Lead SOT-23 package ■ Uses tiny capacitors and inductors ■ Cycle-by-cycle current limiting ■ Internally compensated Applications ■ White LED Current Source ■ PDA’s and Palm-Top Computers ■ Digital Cameras ■ Portable Phones and Games ■ Local Boost Regulator Typical Application Circuit 20059110 20059130 © 2010 National Semiconductor Corporation 200591 www.national.com LM 2731 0.6/1.6 M Hz Boost Converters W ith 22V Internal FET Sw itch in SO T-23 20059153 20059156 20059155 White LED Flash Application www.national.com 2 LM 27 31 Connection Diagram Top View 20059111 5-Lead SOT-23 Package See NS Package Number MF05A Ordering Information Order Number Package Type Package Drawing Supplied As Package ID LM2731XMF SOT23-5 MF05A 1K Tape and Reel S51A LM2731XMFX 3K Tape and Reel S51A LM2731YMF 1K Tape and Reel S51B LM2731YMFX 3K Tape and Reel S51B Pin Descriptions Pin Name Function 1 SW Drain of the internal FET switch. 2 GND Analog and power ground. 3 FB Feedback point that connects to external resistive divider. 4 SHDN Shutdown control input. Connect to Vin if the feature is not used. 5 VIN Analog and power input. 3 www.national.com LM 2731 Absolute Maximum Ratings (Note 1) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. Storage Temperature Range −65°C to +150°C Operating Junction Temperature Range −40°C to +125°C Lead Temp. (Soldering, 5 sec.) 300°C Power Dissipation (Note 2) Internally Limited FB Pin Voltage −0.4V to +6V SW Pin Voltage −0.4V to +22V Input Supply Voltage −0.4V to +14.5V SHDN Pin Voltage −0.4V to VIN + 0.3V θJ-A (SOT23-5) 265°C/W ESD Rating (Note 3) Human Body Model 2 kV Electrical Characteristics Limits in standard typeface are for TJ = 25°C, and limits in boldface type apply over the full operating temperature range (−40°C ≤ TJ ≤ +125°C). Unless otherwise specified: VIN = 5V, VSHDN = 5V, IL = 0A. Symbol Parameter Conditions Min(Note 4) Typical (Note 5) Max (Note 4) Units VIN Input Voltage 2.7 14 V VOUT (MIN) Minimum Output Voltage Under Load RL = 43Ω X Option (Note 8) VIN = 2.7V 5.4 7 V VIN = 3.3V 8 10 VIN = 5V 13 17 RL = 43Ω Y Option (Note 8) VIN = 2.7V 8.25 10 VIN = 3.3V 10.5 12 VIN = 5V 14 16 RL = 15Ω X Option (Note 8) VIN = 2.7V 3.75 5 VIN = 3.3V 5 6.5 VIN = 5V 8.75 11 RL = 15Ω Y Option (Note 8) VIN = 2.7V 5 6 VIN = 3.3V 5.5 7.5 VIN = 5V 9 11 ISW Switch Current Limit (Note 6) 1.8 1.4 2 A RDS(ON) Switch ON Resistance ISW = 100 mA Vin = 5V 260 400 500 mΩ ISW = 100 mA Vin = 3.3V 300 450 550 SHDNTH Shutdown Threshold Device ON 1.5 V Device OFF 0.50 ISHDN Shutdown Pin Bias Current VSHDN = 0 0 µA VSHDN = 5V 0 2 VFB Feedback Pin Reference Voltage VIN = 3V 1.205 1.230 1.255 V IFB Feedback Pin Bias Current VFB = 1.23V 60 500 nA IQ Quiescent Current VSHDN = 5V, Switching "X" 2 3.0 mA VSHDN = 5V, Switching "Y" 1.0 2 VSHDN = 5V, Not Switching 400 500 µA VSHDN = 0 0.024 1 FB Voltage Line Regulation 2.7V ≤ VIN ≤ 14V 0.02 %/V FSW Switching Frequency (Note 7) “X” Option 1 1.6 1.85 MHz“Y” Option 0.40 0.60 0.8 www.national.com 4 LM 27 31 Symbol Parameter Conditions Min(Note 4) Typical (Note 5) Max (Note 4) Units DMAX Maximum Duty Cycle (Note 7) “X” Option 86 93 %“Y” Option 92 96 IL Switch Leakage Not Switching VSW = 5V 1 µA Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating the device outside of the limits set forth under the operating ratings which specify the intended range of operating conditions. Note 2: The maximum power dissipation which can be safely dissipated for any application is a function of the maximum junction temperature, TJ(MAX) = 125° C, the junction-to-ambient thermal resistance for the SOT-23 package, θJ-A = 265°C/W, and the ambient temperature, TA. The maximum allowable power dissipation at any ambient temperature for designs using this device can be calculated using the formula: If power dissipation exceeds the maximum specified above, the internal thermal protection circuitry will protect the device by reducing the output voltage as required to maintain a safe junction temperature. Note 3: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. Note 4: Limits are guaranteed by testing, statistical correlation, or design. Note 5: Typical values are derived from the mean value of a large quantity of samples tested during characterization and represent the most likely expected value of the parameter at room temperature. Note 6: Switch current limit is dependent on duty cycle (see Typical Performance Characteristics). Note 7: Guaranteed limits are the same for Vin = 3.3V input. Note 8: L = 10 µH, COUT = 4.7 µF, duty cycle = maximum Typical Performance Characteristics Unless otherwise specified: VIN = 5V, SHDN pin tied to VIN. Iq Vin (Active) vs Temperature - "X" 20059102 Iq Vin (Active) vs Temperature - "Y" 20059104 5 www.national.com LM 2731 Oscillator Frequency vs Temperature - "X" 20059105 Oscillator Frequency vs Temperature - "Y" 20059101 Max. Duty Cycle vs Temperature - "X" 20059107 Max. Duty Cycle vs Temperature - "Y" 20059106 Iq Vin (Idle) vs Temperature 20059125 Feedback Bias Current vs Temperature 20059126 www.national.com 6 LM 27 31 Feedback Voltage vs Temperature 20059127 RDS(ON) vs Temperature 20059128 Current Limit vs Temperature 20059129 RDS(ON) vs VIN 20059152 Efficiency vs Load Current - "X" VIN = 2.7V, VOUT = 5V 20059135 Efficiency vs Load Current - "X" VIN = 3.3V, VOUT = 5V 20059136 7 www.national.com LM 2731 Efficiency vs Load Current - "X" VIN = 4.2V, VOUT = 5V 20059137 Efficiency vs Load Current - "X" VIN = 2.7V, VOUT = 12V 20059138 Efficiency vs Load Current - "X" VIN = 3.3V, VOUT = 12V 20059139 Efficiency vs Load Current - "X" VIN = 5V, VOUT = 12V 20059140 Efficiency vs Load Current - "X" VIN = 5V, VOUT = 18V 20059141 Efficiency vs Load Current - "Y" VIN = 2.7V, VOUT = 5V 20059142 www.national.com 8 LM 27 31 Efficiency vs Load Current - "Y" VIN = 3.3V, VOUT = 5V 20059143 Efficiency vs Load Current - "Y" VIN = 4.2V, VOUT = 5V 20059144 Efficiency vs Load Current - "Y" VIN = 2.7V, VOUT = 12V 20059145 Efficiency vs Load Current - "Y" VIN = 3.3V, VOUT = 12V 20059146 Efficiency vs Load Current - "Y" VIN = 5V, VOUT = 12V 20059147 9 www.national.com LM 2731 Block Diagram 20059112 Theory of Operation The LM2731 is a switching converter IC that operates at a fixed frequency (0.6 or 1.6 MHz) for fast transient response over a wide input voltage range and incorporates pulse-by- pulse current limiting protection. Because this is current mode control, a 33 mΩ sense resistor in series with the switch FET is used to provide a voltage (which is proportional to the FET current) to both the input of the pulse width modulation (PWM) comparator and the current limit amplifier. At the beginning of each cycle, the S-R latch turns on the FET. As the current through the FET increases, a voltage (propor- tional to this current) is summed with the ramp coming from the ramp generator and then fed into the input of the PWM comparator. When this voltage exceeds the voltage on the other input (coming from the Gm amplifier), the latch resets and turns the FET off. Since the signal coming from the Gm amplifier is derived from the feedback (which samples the voltage at the output), the action of the PWM comparator constantly sets the correct peak current through the FET to keep the output voltage in regulation. Q1 and Q2 along with R3 - R6 form a bandgap voltage refer- ence used by the IC to hold the output in regulation. The currents flowing through Q1 and Q2 will be equal, and the feedback loop will adjust the regulated output to maintain this. Because of this, the regulated output is always maintained at a voltage level equal to the voltage at the FB node "multiplied up" by the ratio of the output resistive divider. The current limit comparator feeds directly into the flip-flop that drives the switch FET. If the FET current reaches the limit threshold, the FET is turned off and the cycle terminated until the next clock pulse. The current limit input terminates the pulse regardless of the status of the output of the PWM com- parator. Application Hints SELECTING THE EXTERNAL CAPACITORS The best capacitors for use with the LM2731 are multi-layer ceramic capacitors. They have the lowest ESR (equivalent series resistance) and highest resonance frequency which makes them optimum for use with high frequency switching converters. When selecting a ceramic capacitor, only X5R and X7R di- electric types should be used. Other types such as Z5U and Y5F have such severe loss of capacitance due to effects of temperature variation and applied voltage, they may provide as little as 20% of rated capacitance in many typical applica- tions. Always consult capacitor manufacturer’s data curves before selecting a capacitor. High-quality ceramic capacitors can be obtained from Taiyo-Yuden, AVX, and Murata. SELECTING THE OUTPUT CAPACITOR A single ceramic capacitor of value 4.7 µF to 10 µF will provide sufficient output capacitance for most applications. If larger amounts of capacitance are desired for improved line support and transient response, tantalum capacitors can be used. Aluminum electrolytics with ultra low ESR such as Sanyo Os- con can be used, but are usually prohibitively expensive. Typical AI electrolytic capacitors are not suitable for switching frequencies above 500 kHz due to significant ringing and temperature rise due to self-heating from ripple current. An output capacitor with excessive ESR can also reduce phase margin and cause instability. In general, if electrolytics are used, it is recommended that they be paralleled with ceramic capacitors to reduce ringing, switching losses, and output voltage ripple. SELECTING THE INPUT CAPACITOR An input capacitor is required to serve as an energy reservoir for the current which must flow into the coil each time the switch turns ON. This capacitor must have extremely low ESR, so ceramic is the best choice. We recommend a nomi- nal value of 2.2 µF, but larger values can be used. Since this www.national.com 10 LM 27 31 capacitor reduces the amount of voltage ripple seen at the input pin, it also reduces the amount of EMI passed back along that line to other circuitry. FEED-FORWARD COMPENSATION Although internally compensated, the feed-forward capacitor Cf is required for stability (see Basic Application Circuit). Adding this capacitor puts a zero in the loop response of the converter. The recommended frequency for the zero fz should be approximately 6 kHz. Cf can be calculated using the for- mula: Cf = 1 / (2 X π X R1 X fz) SELECTING DIODES The external diode used in the typical application should be a Schottky diode. A 20V diode such as the MBR0520 is rec- ommended. The MBR05XX series of diodes are designed to handle a maximum average current of 0.5A. For applications exceed- ing 0.5A average but less than 1A, a Microsemi UPS5817 can be used. LAYOUT HINTS High frequency switching regulators require very careful lay- out of components in order to get stable operation and low noise. All components must be as close as possible to the LM2731 device. It is recommended that a 4-layer PCB be used so that internal ground planes are available. As an example, a recommended layout of components is shown: 20059116 Recommended PCB Component Layout Some additional guidelines to be observed: 1. Keep the path between L1, D1, and C2 extremely short. Parasitic trace inductance in series with D1 and C2 will increase noise and ringing. 2. The feedback components R1, R2 and CF must be kept close to the FB pin of U1 to prevent noise injection on the FB pin trace. 3. If internal ground planes are available (recommended) use vias to connect directly to ground at pin 2 of U1, as well as the negative sides of capacitors C1 and C2. SETTING THE OUTPUT VOLTAGE The output voltage is set using the external resistors R1 and R2 (see Basic Application Circuit). A value of approximately 13.3 kΩ is recommended for R2 to establish a divider current of approximately 92 µA. R1 is calculated using the formula: R1 = R2 X (VOUT/1.23 − 1) SWITCHING FREQUENCY The LM2731 is provided with two switching frequencies: the “X” version is typically 1.6 MHz, while the “Y” version is typi- cally 600 kHz. The best frequency for a specific application must be determined based on the trade-offs involved: Higher switching frequency means the inductors and capac- itors can be made smaller and cheaper for a given output voltage and current. The down side is that efficiency is slightly lower because the fixed switching losses occur more fre- quently and become a larger percentage of total power loss. EMI is typically worse at higher switching frequencies be- cause more EMI energy will be seen in the higher frequency spectrum where most circuits are more sensitive to such in- terference. 11 www.national.com LM 2731 20059117 Basic Application Circuit DUTY CYCLE The maximum duty cycle of the switching regulator deter- mines the maximum boost ratio of output-to-input voltage that the converter can attain in continuous mode of operation. The duty cycle for a given boost application is defined as: This applies for continuous mode operation. INDUCTANCE VALUE The first question we are usually asked is: “How small can I make the inductor?” (because they are the largest sized com- ponent and usually the most costly). The answer is not simple and involves trade-offs in performance. Larger inductors mean less inductor ripple current, which typically means less output voltage ripple (for a given size of output capacitor). Larger inductors also mean more load power can be delivered because the energy stored during each switching cycle is: E = L/2 X (lp)2 Where “lp” is the peak inductor current. An important point to observe is that the LM2731 will limit its switch current based on peak current. This means that since lp(max) is fixed, in- creasing L will increase the maximum amount of power avail- able to the load. Conversely, using too little inductance may limit the amount of load current which can be drawn from the output. Best performance is usually obtained when the converter is operated in “continuous” mode at the load current range of interest, typically giving better load regulation and less output ripple. Continuous operation is defined as not allowing the in- ductor current to drop to zero during the cycle. It should be noted that all boost converters shift over to discontinuous op- eration as the output load is reduced far enough, but a larger inductor stays “continuous” over a wider load current range. To better understand these trade-offs, a typical application circuit (5V to 12V boost with a 10 µH inductor) will be ana- lyzed. We will assume: VIN = 5V, VOUT = 12V, VDIODE = 0.5V, VSW = 0.5V Since the frequency is 1.6 MHz (nominal), the period is ap- proximately 0.625 µs. The duty cycle will be 62.5%, which means the ON time of the switch is 0.390 µs. It should be noted that when the switch is ON, the voltage across the in- ductor is approximately 4.5V. Using the equation: V = L (di/dt) We can then calculate the di/dt rate of the inductor which is found to be 0.45 A/µs during the ON time. Using these facts, we can then show what the inductor current will look like dur- ing operation: 20059118 10 µH Inductor Current, 5V–12V Boost (LM2731X) During the 0.390 µs ON time, the inductor current ramps up 0.176A and ramps down an equal amount during the OFF time. This is defined as the inductor “ripple current”. It can also be seen that if the load current drops to about 33 mA, the inductor current will begin touching the zero axis which means it will be in discontinuous mode. A similar analysis can be performed on any boost converter, to make sure the ripple current is reasonable and continuous operation will be main- tained at the typical load current values. MAXIMUM SWITCH CURRENT The maximum FET switch current available before the current limiter cuts in is dependent on duty cycle of the application. This is illustrated in the graphs below which show typical val- ues of switch current for both the "X" and "Y" versions as a function of effective (actual) duty cycle: www.national.com 12 LM 27 31 20059150 Switch Current Limit vs Duty Cycle - "X" 20059151 Switch Current Limit vs Duty Cycle - "Y" CALCULATING LOAD CURRENT As shown in the figure which depicts inductor current, the load current is related to the average inductor current by the rela- tion: ILOAD = IIND(AVG) x (1 - DC) Where "DC" is the duty cycle of the application. The switch current can be found by: ISW = IIND(AVG) + ½ (IRIPPLE) Inductor ripple current is dependent on inductance, duty cy- cle, input voltage and frequency: IRIPPLE = DC x (VIN-VSW) / (f x L) combining all terms, we can develop an expression which al- lows the maximum available load current to be calculated: The equation shown to calculate maximum load current takes into account the losses in the inductor or turn-OFF switching losses of the FET and diode. For actual load current in typical applications, we took bench data for various input and output voltages for both the "X" and "Y" versions of the LM2731 and displayed the maximum load current available for a typical device in graph form: 20059148 Max. Load Current (typ) vs VIN - "X" 20059149 Max. Load Current (typ) vs VIN - "Y" DESIGN PARAMETERS VSW AND ISW The value of the FET "ON" voltage (referred to as VSW in the equations) is dependent on load current. A good approxima- tion can be obtained by multiplying the "ON Resistance" of the FET times the average inductor current. FET on resistance increases at VIN values below 5V, since the internal N-FET has less gate voltage in this input voltage range (see Typical performance Characteristics curves). Above VIN = 5V, the FET gate voltage is internally clamped to 5V. The maximum peak switch current the device can deliver is dependent on duty cycle. For higher duty cycles, see Typical performance Characteristics curves. THERMAL CONSIDERATIONS At higher duty cycles, the increased ON time of the FET means the maximum output current will be determined by power dissipation within the LM2731 FET switch. The switch power dissipation from ON-state conduction is calculated by: P(SW) = DC x IIND(AVE)2 x RDS(ON) 13 www.national.com LM 2731 There will be some switching losses as well, so some derating needs to be applied when calculating IC power dissipation. INDUCTOR SUPPLIERS Recommended suppliers of inductors for this product include, but are not limited to Sumida, Coilcraft, Panaso